This invention relates to a series resonant converter, which comprises a series resonant circuit, a rectifier in series therewith and a semiconductor switch and, the series resonant converter being operative to convert a DC voltage into a different DC voltage supplied to a load.
A series resonant converter usually has two resonant loops each of which comprises a series resonant circuit consisting of an inductor and a capacitor, a rectifier connected in series with the series resonant circuit and a switch element, e.g., a bipolar transistor (or MOS transistor) or a diode. The resonant current flowing through the switch element has a sinusoidal waveform which naturally goes through zero, so that the switch element need not forcibly turn the current off. Theoretically, therefore, there is no switching loss. Thus, less noise is produced, and it is possible to readily construct the converter so that it is operable at a high frequency. In addition, it can be expected to obtain a noise-free, small size and light weight converter. Further, the output characteristics of the series resonant circuit are essentially constant current characteristics. Therefore, the converter system can be readily protected when there occurs an overload or a short-circuit in the load.
FIG. 1 shows a prior art series resonant converter which is disclosed in, for instance, W. Mc Murray, "The Thyristor Electronic Transformer: A Power Converter Using a High Frequency Link", IEEE Transaction on IGA, No. 4, PP. 451-457. FIGS. 2A, 2B and 2C are waveform diagrams illustrating the operation of the converter.
Referring to FIG. 1, a reference numeral 11 designates a semiconductor switch, which is a transistor and serves as a switch element. When semiconductor switch 11 is turned on by application of a gate signal to the gate of transistor 11, series resonant current i.sub.1 (shown by solid line) is caused to flow through a series resonant loop extending from power supply 12 through semiconductor switch 11, rectifier 13, load 14 (capacitor 15), then rectifier 13 again, then capacitor 16 and inductor 17 to return to power supply 12. This series resonant current i.sub.1 is sinusoidal as shown in FIG. 2A. Denoting the inductance of inductor 17 by L.sub.0, the capacitance of capacitor 16 by C.sub.0 and the capacitance of capacitor 15 by C.sub.0 t, since C.sub.0 t&gt;&gt;C.sub.0, the resonant current i.sub.1 becomes zero at a time .pi..sqroot.L.sub.0 C.sub.0 after it has been caused to flow. At that time the voltage V.sub.1 across capacitor 16 assumes a peak voltage V.sub.12. When the peak voltage V.sub.12 is higher than the sum of output voltage V.sub.0 (i.e., the voltage across load 14) and voltage E/2 of power supply 12 at this time, as shown in FIG. 2C, a reverse curent i.sub.i ' as shown by dashed line in FIG. 2A is caused to flow through diode 21. This current i.sub.i' again is a series resonant current through inductor 17 and capacitor 16 and after a period .pi..sqroot.L.sub.0 C.sub.0 from its state, turns to zero. The gate signal is arranged to have a duration longer than .pi..sqroot.L.sub.0 C.sub.0 and shorter than 2.pi..sqroot.L.sub.0 C.sub.0. Therefore, the gate signal having been applied to transistor 11 is removed within the period of current i.sub.1 ', preventing generation of further resonant current after the cessation of i.sub.1 '. The currents i.sub.1 and i.sub.1 ' are full-wave rectified through rectifier 13 so that current i.sub.3 flows through capacitor 15 for filtering as shown in FIG. 2B. Capacitor 15 continuously supplies a DC voltage across load 14.
Reference numeral 18 designates another semiconductor switch, which also is a transistor and serves as a switch element. When semiconductor switch 18 is turned on, series resonant current i.sub.2 (shown by solid line) is caused to flow through a series resonant loop extending from power supply 19 through inductor 17, capacitor 16, rectifier 13, load 14 (capacitor 15), then rectifier 13 again, and then semiconductor switch 18 to return to power supply 19. This series resonant current i.sub.2 is shown in FIG. 2A. After i.sub.2 has once become zero, current i.sub.2 ' flows through diode 22. Rectifier 13 rectifies these currents i.sub.2 and i.sub.2 ' to provide current i.sub.3 (FIG. 2B) for charging capacitor 15 for filtering. That is, capacitor 15 supplies a DC voltage to load 14. In the above operation currents i.sub.1 and i.sub.2 are series resonant currents equal in magnitude and opposite in polarity to each other. Similarly, series resonant currents i.sub.1 ' and i.sub.2 ' are equal in magnitude and opposite in polarity to each other.
Since the currents which are caused to flow through semiconductor switches 11 and 18 are series resonant currents, they extinguish themselves to zero a time .pi..sqroot.L.sub.0 C.sub.0 after the semiconductor switches 11 and 18 have been turned on. Therefore, there is no need to forcibly switch off the currents through semiconductor switches 11 and 18, that is, there is inherently no switching loss. It is thus possible to obtain a high frequency operation.
In this prior art series resonant converter, the quantity of charge transmitted to the load through rectifier 13 can be calculated from changes in the voltage across capacitor 16. Referring to FIG. 2C which shows the waveform of the voltage V.sub.1 across capacitor 16, normally .vertline.V.sub.11 .vertline.=.vertline.V.sub.11 '.vertline., where V.sub.11 is the voltage across capacitor 16 before the resonant current is caused to flow through one of switches 11 and 18, and V.sub.11 ' is the voltage across capacitor before the next resonant current is caused to flow through the other switch. Quantity Q.sub.0 of charge that is transferred to the load per half cycle period is thus EQU Q.sub.0 =C.sub.0 {(V.sub.11 +V.sub.12)+(V.sub.12 -V.sub.11 ')}=2C.sub.0 V.sub.12 ( 1)
where V.sub.12 is the peak voltage across capacitor 16. Average value I.sub.3 of current i.sub.3 shown in FIG. 2B is EQU I.sub.3 =Q/(1/2f)=4C.sub.0 V.sub.12 f (2)
where f is the operating frequency, at which semiconductor switches 11 and 18 are alternately turned on. Normally, the voltage across filter capacitor 15 is fixed, so that current I.sub.3 is entirely supplied to load 14. Output voltage V.sub.0 is thus EQU V.sub.0 =R.multidot.I.sub.3 =(4C.sub.0 V.sub.12 f).multidot.R (3)
where R is the resistance of load 14.
From equation (3) it will be seen that output DC voltage V.sub.0 can be controlled through control of either C.sub.0, V.sub.12 or f. At present, it is difficult to continuously control C.sub.0. Peak voltage V.sub.12 is normally clamped by power sources 12 and 19 to E/2 in stationary states for there are feedback currents i.sub.1 ' and i.sub.2 ' through diode 21 or 22 in parallel with semiconductor switch 11 or 18. Voltage V.sub.12, therefore, can not be controlled unless the voltages of power sources 12 and 19 are varied. Usually, therefore, the voltage control of V.sub.0 to a constant voltage is effected through control of operating frequency f.
Although not shown in the drawings, it is conventionally arranged that a gate signal generating circuit generates two gate signals of a frequency f and 180.degree. out of phase from each other which are supplied to the respective gates of switches 11 and 18, and the output voltage V.sub.0 is fed back to the signal generating circuit to control the frequency f of the gate signals so as to maintain the output voltage V.sub.0 constant. Therefore, frequency f lowers as resistance R increases, as will be seen from equation (3), and in the case of light load operation, the frequency may easily enter an audio frequency range. For example, if operating frequency f is 100 kHz under the rated current (100% load) condition, a load reduction to 20% load or less would cause the frequency f to become lower than 20 kHz, giving rise to noisy sounds. It has been difficult to cope with the noise because the operating frequency changes with the load.
FIG. 3 shows a series resonant converter proposed in Electronic Engineering, September 1981, page 39 to solve the problems discussed above. In this proposed series resonant converter, a series circuit consisting of the primary winding of transformer 27 and capacitor 16 is connected between the junction of switch elements 11 and 18 and one terminal of inductor 17. DC power sources 12 and 19 are connected between the other terminal of inductor 17 and the other terminals of respective switch elements 11 and 18. Capacitor 44 is connected in parallel with the secondary winding of transformer 17, and rectifier 13, inductor 20 and load 14 are connected in series with the secondary winding of transformer 27. A feature of this prior art converter resides in that the switching frequency, at which switch elements 11 and 18 are turned on and off, is set variable in a higher frequency range than resonant frequency f.sub.0 =1/(2.pi..sqroot.L.sub.0 C.sub.0) based on inductance L.sub.0 of inductance 17 and capacitance C.sub.0 of capacitor 16. The current in the series resonant circuit at a frequency higher than f.sub.0 is inversely proportional to the frequency. Therefore, it is possible to obtain constant voltage control through switching frequency control such that the switching frequency is increased with reducing load current and reduced with increasing load current in a range above f.sub.0. Thus, it is possible to eliminate noise by selecting f.sub.0 to be above the audible frequency range. However, since the switching frequency is above f.sub.0, switch element 11 or 18 can not be made to turn on after the resonant current through inductor 17 has become zero. In other words, a suddenly increasing switching current is caused when the switch element is turned on, leading to switching loss increase and noise increase. To reduce noise and also further reduce the switching frequency variation range in this prior art converter, capacitor 44 is connected across the secondary winding of transformer 27, and series inductor 20 is connected between the output of rectifier 13 and one terminal of capacitor 15, whereby a sinusoidal voltage is transmitted to the secondary side of transformer 27. However, a very large reactive current flows through capacitor 44 without being substantially influenced by the load. In other words, a substantial portion of the current flowing through switch elements 11 and 18 under a light load condition is a reactive current flowing through transformer 27 to capacitor 44, so that the efficiency is extremely reduced.
FIG. 4 shows a further well-known series resonant converter. In this converter diodes 21 and 22 are connected in parallel with respective capacitors 41 and 42, DC power supply 43 is applied across the series connection of capacitors 41 and 42, and rectifier 13 is serially connected between the junction of semiconductor switches 11 and 18 and inductor 17. The gates of transistors (i.e. switch elements) 11 and 18 are controlled by gate signals having a frequency and a 50% duty ratio, 180.degree. out of phase from each other. This converter has a feature that the endurable voltage (i.e. breakdown voltage) required for semiconductor switches 11 and 18 and resonant capacitors 41 and 42 may suffice if selected to be higher than the voltage across power supply 43. Moreover, since the effective value of resonant current is small compared to that of the series resonant converter shown in FIG. 1, the loss at switches 11, 18 and rectifier 13 is small and also no switching loss will be caused in contrast with the case of FIG. 3.
The operation of this converter will now be described under the assumption of an initial condition wherein resonant capacitors 41 and 42 have been charged respectively to the voltage of power supply 43 and to zero voltage. FIG. 5 shows waveforms obtained in various parts of the converter of FIG. 4. When semiconductor switch 11 is turned on, charging current i.sub.1 flows from power supply 43 through semiconductor switch 11, rectifier 13, load 14 (capacitor 15), then rectifier 13 again, and then resonant inductor 17 to charge resonant capacitor 42. At the same time, discharging current i.sub.2 from resonant capacitor 41 flows through semiconductor switch 11, rectifier 13, load 14 (capacitor 15), then rectifier 13 again, and then resonant capacitor 17. Currents i.sub.1 and i.sub.2 are equal and assume a sinusoidal waveform with a half cycle period .pi..sqroot.2L.sub.0 C.sub.0 as shown in FIG. 5A. They are resonant currents which charge and discharge resonant capacitor 42 as shown in FIG. 5D and discharge and charge resonant capacitor 41 as shown in FIG. 5C. Denoting the voltage across capacitor 15 (i.e., the output voltage) by V.sub. 0, the inductance of resonant inductor 17 by L.sub.0, the capacitance of resonant capacitors 41 and 42 by C.sub.0 and the voltage of power supply 12 by E, then the voltages Vc.sub.1 and Vc.sub.2 across resonant capacitors 41 and 42 become respectively zero and E at t.sub.1 after the lapse of time of approximately ##EQU1## from time t.sub.o. At this moment t.sub.1, diode 21 is turned on, so that the current which has been flowing through resonant inductor 17 now turns to flow as current i.sub.2 ' shown in FIG. 5B through resonant inductor 17, diode 21, semiconductor switch 11, rectifier 13, load 14 (capacitor 15) and then rectifier 13 again, as shown in FIG. 4. This current i.sub.2 ' becomes zero after the lapse of time from the instant of turning-on of diode 21, thus entering into a cease period which continues until semiconductor switch 11 turns off and switch 18 turns on at t.sub.3 to end one half cycle of operation. In the next half cycle, semiconductor switch 18 is turned on at t.sub.3 to charge resonant capacitor 41 and discharge resonant capacitor 42.
With the prior art series resonant converter shown in FIG. 1, the output voltage V.sub.0 versus output current I.sub.0 characteristic is a constant current characteristic as shown in FIG. 6A; that is, output current I.sub.0 is substantially constant regardless of reduction of output voltage V.sub.0 so long as the operating frequency f is constant. With the series resonant converter shown in FIG. 4, the average of currents i.sub.1, i.sub.2 ' flowing into rectifier 13, i.e. output current I.sub.0, is proportional to capacitance C.sub.0, operating frequency f and inversely proportional to output voltage V.sub.0 as shown in FIG. 6B. Therefore, when V.sub.0 is extraordinarily reduced due to such cases as an output short-circuit while operating frequency f is held clamped, T.sub.2 (see FIG. 5B) is increased extremely in accordance with equation (5). In this case, the currents through semiconductor switches 11 and 18 no longer extinguish themselves within each half cycle of the operating frequency f, so that it is necessary to forcibly turn off these currents by semiconductor switches 11 and 18, thus leading to problems of noise increase and switching loss increase. When output voltage V.sub.0 becomes low because of heavy load, for example, it is necessary to reduce operating frequency f to hold output current I.sub.0 within maximum permissible load current Im shown by dashed line 45. Depending on the load condition, the operation frequency enters the audible frequency range f.sub.A, thus giving rise to noise.
A primary object of the invention is to provide a series resonant converter which permits to set a lowest allowable frequency to change in the operating frequency relative to load variations and which is free from noise.
A second object of the invention is to provide a series resonant converter which permits constant voltage control for over a range from zero to full load and has a narrow operating frequency range as well as capability of holding the operating frequency above a lowest allowable frequency.
A third object of the invention is to provide a series resonant circuit which can limit the output voltage without noise generation even when the output voltage is extraordinarily reduced due to such causes as an output short-circuit.